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1 LT1370 500khz high efficiency 6a switching regulator the lt ? 1370 is a monolithic high frequency current mode switching regulator. it can be operated in all standard switching configurations including boost, buck, flyback, forward, inverting and cuk. a 6a high efficiency switch is included on the die, along with all oscillator, control and protection circuitry. the LT1370 typically consumes only 4.5ma quiescent current and has higher efficiency than previous parts. high frequency switching allows for very small inductors to be used. new design techniques increase flexibility and maintain ease of use. switching is easily synchronized to an exter- nal logic level source. a logic low on the shutdown pin reduces supply current to 12 m a. unique error amplifier circuitry can regulate positive or negative output voltage while maintaining simple frequency compensation tech- niques. nonlinear error amplifier transconductance reduces output overshoot on start-up or overload recov- ery. oscillator frequency shifting protects external com- ponents during overload conditions. descriptio n u n faster switching with increased efficiency n uses small inductors: 4.7 m h n all surface mount components n low minimum supply voltage: 2.7v n quiescent current: 4.5ma typ n current limited power switch: 6a n regulates positive or negative outputs n shutdown supply current: 12 m a typ n easy external synchronization n switch resistance: 0.065 w typ typical applicatio n u features , ltc and lt are registered trademarks of linear technology corporation. n boost regulators n laptop computer supplies n multiple output flyback supplies n inverting supplies applicatio n s u 5v to 12v boost converter LT1370 v in v c 5v gnd fb LT1370 ?ta01 v sw s/s l1* c1** 22 m f 25v c4** 22 f 25v 2 c2 0.047 m f c3 0.0047 m f r3 2k r2 6.19k 1% r1 53.6k 1% v out ? 12v d1 mbrd835l on off * ** coiltronics up2-4r7 (4.7 h) up4-100 (10 h) avx tpsd226m025r0200 + + l1 4.7 m h 10 m h i out 1.8a 2a ? max i out 12v output efficiency load current (a) 0 80 efficiency (%) 84 82 86 88 90 1.0 1.2 1.4 1.6 1.8 0.2 0.4 0.6 0.8 2.0 LT1370 ?ta02 92 v in = 5v l = 10 m h
2 LT1370 a u g w a w u w a r b s o lu t exi t i s supply voltage ....................................................... 30v switch voltage LT1370 ............................................................... 35v LT1370hv .......................................................... 42v s/s, shdn, sync pin voltage ................................ 30v feedback pin voltage (transient, 10ms) .............. 10v feedback pin current ........................................... 10ma negative feedback pin voltage (transient, 10ms) ............................................. 10v operating ambient temperature range ...... 0 c to 70 c operating junction temperature range commercial .......................................... 0 c to 125 c industrial ......................................... C 40 c to 125 c storage temperature range ................ C 65 c to 150 c lead temperature (soldering, 10 sec)................. 300 c e lectr ic al c c hara terist ics v in = 5v, v c = 0.6v, v fb = v ref , v sw , s/s and nfb pins open, t a = 25 c unless otherwise noted. symbol parameter conditions min typ max units v ref reference voltage measured at feedback pin 1.230 1.245 1.260 v v c = 0.8v l 1.225 1.245 1.265 v i fb feedback input current v fb = v ref 250 550 na l 900 na reference voltage line regulation 2.7v v in 25v, v c = 0.8v l 0.01 0.03 %/v v nfr negative feedback reference voltage measured at negative feedback pin C 2.525 C 2.48 C 2.435 v feedback pin open, v c = 0.8v l C 2.560 C 2.48 C 2.400 v i nfb negative feedback input current v nfb = v nfr l C45 C30 C15 m a negative feedback reference voltage 2.7v v in 25v, v c = 0.8v l 0.01 0.05 %/v line regulation g m error amplifier transconductance d i c = 25 m a 1100 1500 1900 m mho l 700 2300 m mho error amplifier source current v fb = v ref C 150mv, v c = 1.5v l 120 200 350 m a error amplifier sink current v fb = v ref + 150mv, v c = 1.5v l 1400 2400 m a wu u package / o rder i for atio t jmax = 125 c, q ja = 30 c/w, q jc = 4 c/w r package 7-lead plastic dd front view tab is gnd v in nfb v sw gnd s/s fb v c 7 6 5 4 3 2 1 order part number with package soldered to 0.5 inch 2 copper area over backside ground plane or internal power plane. q ja can vary from 20 c/w to >40 c/w depending on mounting technique order part number t7 package 7-lead to-220 v in nfb v sw gnd s/s fb v c front view 7 6 5 4 3 2 1 tab is gnd t jmax = 125 c, q ja = 50 c/w, q jc = 4 c/w consult factory for military grade parts. LT1370cr LT1370hvcr LT1370ir LT1370hvir LT1370ct7 LT1370hvct7 LT1370it7 LT1370hvit7
3 LT1370 symbol parameter conditions min typ max units error amplifier clamp voltage high clamp, v fb = 1v 1.5 1.8 2.30 v low clamp, v fb = 1.5v 0.2 0.3 0.52 v a v error amplifier voltage gain 500 v/ v v c pin threshold duty cycle = 0% 0.9 1.1 1.35 v f switching frequency 2.7v v in 25v 460 500 550 khz 0 c t j 125 c l 440 500 580 khz C40 c t j 0 c (i-grade) 400 580 khz maximum switch duty cycle l 85 95 % switch current limit blanking time 130 300 ns bv output switch breakdown voltage LT1370 l 35 44 v LT1370hvc, 0 c t j 125 c l 42 47 v LT1370hvi, C 40 c t j 0 c (i-grade) 40 v v sat output switch on resistance i sw = 6a l 0.065 0.11 w i lim switch current limit duty cycle = 50% l 6 8 10 a duty cycle = 80% (note 1) 7 a d i in supply current increase during switch on time 22 33 ma/a d i sw control voltage to switch current 10 a/v transconductance minimum input voltage l 2.4 2.7 v i q supply current 2.7v v in 25v l 4.5 6 ma shutdown supply current 2.7v v in 25v, v s/s 0.6v l 12 40 m a shutdown threshold 2.7v v in 25v l 0.6 1.3 2 v shutdown delay l 41225 m s s/s input current 0v s/s 5v l C7 10 m a synchronization frequency range l 600 800 khz e lectr ic al c c hara terist ics v in = 5v, v c = 0.6v, v fb = v ref , v sw , s/s and nfb pins open, t a = 25 c unless otherwise noted. the l denotes specifications which apply over the full operating temperature range. note 1: for duty cycles (dc) between 45% and 85%, minimum switch current limit is given by i lim = 2.65(2.7 C dc).
4 LT1370 typical perfor m a n ce characteristics u w switching frequency vs feedback pin voltage voltage (v) ? input current ( m a) 1 7 LT1370 ?g07 ? ? 0 2 ? ? 1 3 5 0 2 4 6 feedback pin voltage (v) 0 switching frequency (% of typical) 70 90 110 0.8 LT1370 ?g08 50 30 60 80 100 40 20 10 0.2 0.4 0.6 0.1 0.9 0.3 0.5 0.7 1.0 switch saturation voltage vs switch current duty cycle (%) 6.6 switch current limit (a) 7.4 7.2 7.8 8.2 7.0 6.8 7.6 8.0 20 40 60 80 LT1370 ?g02 100 10 0 30 50 70 90 switch current limit vs duty cycle switch current (a) 0 switch voltage (mv) 300 400 550 5 LT1370 ?g01 200 100 250 350 450 500 150 50 0 1 23 4 6 125 c 75 c 25 c 0 c temperature ( c) ?0 1.8 input voltage (v) 2.0 2.2 2.4 2.6 050 100 150 LT1370 ?g03 2.8 3.0 ?5 25 75 125 minimum input voltage vs temperature temperature ( c) ?0 0 shutdown delay ( m s) shutdown threshold (v) 2 6 8 10 20 14 0 50 75 LT1370 ?g04 4 16 18 12 0 0.2 0.6 0.8 1.0 2.0 1.4 0.4 1.6 1.8 1.2 ?5 25 100 125 150 shutdown threshold shutdown delay shutdown delay and threshold vs temperature error amplifier output current vs feedback pin voltage feedback pin voltage (v) 400 error amplifier output current ( m a) 300 200 100 300 100 0.1 0.1 200 0 0.3 0.2 v ref ?5 c 125 c 25 c LT1370 ?g06 temperature ( c) ?0 0 minimum synchronization voltage (v p-p ) 0.5 1.0 1.5 2.0 050 100 150 LT1370 ?g05 2.5 3.0 ?5 25 75 125 f sync = 700khz minimum synchronization voltage vs temperature error amplifier transconductance vs temperature temperature ( c) ?0 0 transconductance ( m mho) 200 600 800 1000 2000 1400 0 50 75 LT1370 ?g09 400 1600 1800 1200 ?5 25 100 125 150 g m = d i (v c ) d v (fb) s/s pin input current vs voltage
5 LT1370 typical perfor m a n ce characteristics u w temperature ( c) ?0 feedback input current (na) 400 500 600 150 LT1370 ?g11 300 200 0 0 50 100 100 800 700 ?5 25 75 125 v fb =v ref feedback input current vs temperature temperature ( c) ?0 ?0 negative feedback input current ( m a) ?0 0 0 50 75 LT1370 ?g12 ?0 ?0 ?0 ?5 25 100 125 150 v nfb =v nfr negative feedback input current vs temperature v c pin threshold and high clamp voltage vs temperature temperature ( c) ?0 1.0 v c voltage (v) 1.4 2.2 0 50 75 LT1370 ?g10 1.2 1.8 2.0 1.6 ?5 25 100 125 150 v c high clamp v c threshold pi n fu n ctio n s uuu v c : the compensation pin is used for frequency compen- sation, current limiting and soft start. it is the output of the error amplifier and the input of the current comparator. loop frequency compensation can be performed with an rc network connected from the v c pin to ground. see applications information. fb: t he feedback pin is used for positive output voltage sensing and oscillator frequency shifting. it is the invert- ing input to the error amplifier. the noninverting input of this amplifier is internally tied to a 1.245v reference. nfb: the negative feedback pin is used for negative output voltage sensing. it is connected to the inverting input of the negative feedback amplifier through a 100k source resistor. s/s: shutdown and synchronization pin. the s/s pin is logic level compatible. shutdown is active low and the shutdown threshold is typically 1.3v. for normal opera- tion, pull the s/s pin high, tie it to v in or leave it floating. to synchronize switching, drive the s/s pin between 600khz and 800khz. see applications information. v in : bypass input supply pin with a low esr capacitor, 10 m f or more. the regulator goes into undervoltage lock- out when v in drops below 2.5v. undervoltage lockout stops switching and pulls the v c pin low. v sw : the switch pin is the collector of the power switch and has large currents flowing through it. keep the traces to the switching components as short as possible to minimize radiation and voltage spikes. gnd: tie all ground pins to a good quality ground plane. see applications information.
6 LT1370 block diagra m w operatio n u the LT1370 is a current mode switcher. this means that switch duty cycle is directly controlled by switch current rather than by output voltage. referring to the block diagram, the switch is turned on at the start of each oscillator cycle. it is turned off when switch current reaches a predetermined level. control of output voltage is obtained by using the output of a voltage sensing error amplifier to set current trip level. this technique has several advantages. first, it has immediate response to input voltage variations, unlike voltage mode switchers which have notoriously poor line transient response. second, it reduces the 90 phase shift at midfrequencies in the energy storage inductor. this greatly simplifies closed-loop frequency compensation under widely vary- ing input voltage or output load conditions. finally, it allows simple pulse-by-pulse current limiting to provide maximum switch protection under output overload or short conditions. a low dropout internal regulator pro- vides a 2.3v supply for all internal circuitry. this low dropout design allows input voltage to vary from 2.7v to 25v with virtually no change in device performance. a 500khz oscillator is the basic clock for all internal timing. it turns on the output switch via the logic and driver circuitry. special adaptive antisat circuitry detects onset of saturation in the power switch and adjusts driver current instantaneously to limit switch saturation. this minimizes driver dissipation and provides very rapid turn-off of the switch. a 1.245v bandgap reference biases the positive input of the error amplifier. the negative input of the amplifier is brought out for positive output voltage sensing. the error amplifier has nonlinear transconductance to reduce out- put overshoot on start-up or overload recovery. when the feedback voltage exceeds the reference by 40mv, error amplifier transconductance increases 10 times, which reduces output overshoot. the feedback input also invokes oscillator frequency shifting, which helps pro- tect components during overload conditions. when the feedback voltage drops below 0.6v, the oscillator fre- quency is reduced 5:1. lower switching frequency allows full control of switch current limit by reducing minimum switch duty cycle. + nfba nfb s/s fb 100k 50k 0.005 w + ea v c v in gnd LT1370 ?bd gnd sense 1.245v ref 5:1 frequency shift osc sync shutdown delay and reset low dropout 2.3v reg anti-sat logic driver sw switch + ia a v ? 20 comp
7 LT1370 unique error amplifier circuitry allows the LT1370 to directly regulate negative output voltages. the negative feedback amplifiers 100k source resistor is brought out for negative output voltage sensing. the nfb pin regulates at C 2.48v while the amplifier output internally drives the fb pin to 1.245v. this architecture, which uses the same main error amplifier, prevents duplicating functions and maintains ease of use. consult ltc marketing for units that can regulate down to C 1.25v. the error signal developed at the amplifier output is brought out externally. this pin (v c ) has three different functions. it is used for frequency compensation, current limit adjustment and soft starting. during normal regula- tor operation this pin sits at a voltage between 1v (low output current) and 1.9v (high output current). the error amplifier is a current output (g m ) type, so this voltage can be externally clamped for lowering current limit. like- wise, a capacitor coupled external clamp will provide soft start. switch duty cycle goes to zero if the v c pin is pulled below the control pin threshold, placing the LT1370 in an idle mode. positive output voltage setting the LT1370 develops a 1.245v reference (v ref ) from the fb pin to ground. output voltage is set by connecting the fb pin to an output resistor divider (figure 1). the fb pin bias current represents a small error and can usually be ignored for values of r2 up to 7k. the suggested value for r2 is 6.19k. the nfb pin is normally left open for positive output applications. positive fixed voltage versions are available (consult ltc marketing). negative output voltage setting the LT1370 develops a C 2.48v reference (v nfr ) from the nfb pin to ground. output voltage is set by connecting the nfb pin to an output resistor divider (figure 2). the C30 m a nfb pin bias current (i nfb ) can cause output voltage errors and should not be ignored. this has been accounted for in the formula in figure 2. the suggested value for r2 is 2.49k. the fb pin is normally left open for negative output applications. dual polarity output voltage sensing certain applications benefit from sensing both positive and negative output voltages. one example is the dual output flyback converter with overvoltage protection circuit shown in the typical applications section. each output voltage resistor divider is individually set as described above. when both the fb and nfb pins are used, r1 v out = v ref 1 + r2 fb pin v ref v out () r1 r2 r1 = r2 ?1 () v out 1.245 LT1370 ?f01 applicatio s i for atio uu w u figure 1. positive output resistor divider the LT1370 acts to prevent either output from going beyond its set output voltage. for example, in this applica- tion if the positive output were more heavily loaded than the negative, the negative output would be greater and would regulate at the desired set-point voltage. the posi- tive output would sag slightly below its set-point voltage. this technique prevents either output from going unregu- lated high at no load. figure 2. negative output resistor divider r1 ? out = v nfb + i nfb (r1) 1 + r2 LT1370 ?f02 nfb pin v nfr i nfb ? out () r1 r2 r1 = + 30 ?10 6 ? v out ? ?2.48 ( ) ( ) 2.48 r2 operatio n u
8 LT1370 shutdown and synchronization the device has a dual function s/s pin which is used for both shutdown and synchronization. this pin is logic level compatible and can be pulled high, tied to v in or left floating for normal operation. a logic low on the s/s pin activates shutdown, reducing the parts supply current to 12 m a. typical synchronization range is from 1.05 to 1.8 times the parts natural switching frequency, but is only guaranteed between 600khz and 800khz. a 12 m s resetable shutdown delay network guarantees the part will not go into shutdown while receiving a synchronization signal. caution should be used when synchronizing above 700khz because at higher sync frequencies the amplitude of the internal slope compensation used to prevent subhar- monic switching is reduced. this type of subharmonic switching only occurs when the duty cycle of the switch is above 50%. higher inductor values will tend to elimi- nate this problem. thermal considerations care should be taken to ensure that the worst-case input voltage and load current conditions do not cause exces- sive die temperatures. typical thermal resistance is 30 c/w for the r package and 50 c/w for the t7 package but these numbers will vary depending on the mounting techniques (copper area, airflow, etc.). heat is transferred from the package via the tab. average supply current (including driver current) is: i in = 4.5ma + dc(i sw /45) i sw = switch current dc = switch duty cycle switch power dissipation is given by: p sw = (i sw ) 2 (r sw )(dc) r sw = output switch on resistance total power dissipation of the die is the sum of supply current times supply voltage, plus switch power: p d(total) = (i in )(v in ) + p sw surface mount heat sinks are available which can lower package thermal resistance by two or three times. one manufacturer, wakefield engineering, offers surface mount heat sinks for the r package and can be reached at (617) 245-5900 or at www.wakefield.com. choosing the inductor for most applications the inductor will fall in the range of 2.2 m h to 22 m h. lower values are chosen to reduce physi- cal size of the inductor. higher values allow more output current because they reduce peak current seen by the power switch, which has a 6a limit. higher values also reduce input ripple voltage and reduce core loss. when choosing an inductor you need to consider maxi- mum load current, core and copper losses, allowable component height, output voltage ripple, emi, fault current in the inductor, saturation and, of course, cost. the following procedure is suggested as a way of handling these somewhat complicated and conflicting requirements. 1. assume that the average inductor current for a boost converter is equal to load current times v out /v in and decide whether or not the inductor must withstand continuous overload conditions. if average inductor current at maximum load current is 3a, for instance, a 3a inductor may not survive a continuous 6a overload condition. also be aware that boost converters are not short-circuit protected and that, under output short conditions, inductor current is limited only by the available current of the input supply. 2. calculate peak inductor current at full load current to ensure that the inductor will not saturate. peak current can be significantly higher than output current, espe- cially with smaller inductors and lighter loads, so dont omit this step. powdered iron cores are forgiving because they saturate softly, whereas ferrite cores saturate abruptly and other core materials fall in between. the following formula assumes continuous mode operation but it errs only slightly on the high side for discontinuous mode, so it can be used for all conditions. applicatio s i for atio uu w u
9 LT1370 applicatio s i for atio uu w u tested for low esr, so they give the lowest esr for a given volume. to further reduce esr, multiple output capaci- tors can be used in parallel. the value in microfarads is not particularly critical, and values from 22 m f to greater than 500 m f work well, but you cannot cheat mother nature on esr. if you find a tiny 22 m f solid tantalum capacitor, it will have high esr and output ripple voltage will be terrible. table 1 shows some typical solid tantalum surface mount capacitors. table 1. surface mount solid tantalum capacitor esr and ripple current e case size esr (max w ) ripple current (a) avx tps, sprague 593d 0.1 to 0.3 0.7 to 1.1 avx taj 0.7 to 0.9 0.4 d case size avx tps, sprague 593d 0.1 to 0.3 0.7 to 1.1 avx taj 0.9 to 2.0 0.36 to 0.24 c case size avx tps 0.2 (typ) 0.5 (typ) avx taj 1.8 to 3.0 0.22 to 0.17 b case size avx taj 2.5 to 10 0.16 to 0.08 many engineers have heard that solid tantalum capacitors are prone to failure if they undergo high surge currents. this is historically true and avx type tps capacitors are specially tested for surge capability, but surge ruggedness is not a critical issue with the output capacitor. solid tantalum capacitors fail during very high turn-on surges, which do not occur at the output of regulators. high discharge surges, such as when the regulator output is dead-shorted, do not harm the capacitors. single inductor boost regulators have large rms ripple current in the output capacitor, which must be rated to handle the current. the formula to calculate this is: output capacitor ripple current (rms) i ripple (rms) = i out = i out v out v in v in dc 1 ?dc dc = switch duty cycle i peak = (i out ) v in = minimum input voltage f = 500khz switching frequency + v out v in v in (v out v in ) 2(f)(l)(v out ) ) ) 3. decide if the design can tolerate an open core geom- etry, like a rod or barrel, which has high magnetic field radiation, or whether it needs a closed core, like a toroid, to prevent emi problems. one would not want an open core next to a magnetic storage media, for instance! this is a tough decision because the rods or barrels are temptingly cheap and small and there are no helpful guidelines to calculate when the magnetic field radiation will be a problem. 4. start shopping for an inductor that meets the requirements of core shape, peak current (to avoid saturation), average current (to limit heating) and fault current. if the inductor gets too hot, wire insulation will melt and cause turn-to-turn shorts. keep in mind that all good things like high efficiency, low profile and high temperature operation will increase cost, sometimes dramatically. 5. after making an initial choice, consider the secondary things like output voltage ripple, second sourcing, etc. use the experts in the ltc applications department if you feel uncertain about the final choice. they have experience with a wide range of inductor types and can tell you about the latest developments in low profile, surface mounting, etc. output capacitor the output capacitor is normally chosen by its effective series resistance (esr), because this is what determines output ripple voltage. at 500khz any polarized capacitor is essentially resistive. to get low esr takes volume , so physically smaller capacitors have high esr. the esr range needed for typical LT1370 applications is 0.025 w to 0.2 w . a typical output capacitor is an avx type tps, 22 m f at 25v (two each), with a guaranteed esr less than 0.2 w . this is a d size surface mount solid tantalum capacitor. tps capacitors are specially constructed and
10 LT1370 applicatio s i for atio uu w u output diode the suggested output diode (d1) is a motorola mbrd835l. it is rated at 8a average forward current and 35v reverse voltage. typical forward voltage is 0.4v at 3a. the diode conducts current only during switch off time. peak re- verse voltage for boost converters is equal to regulator output voltage. average forward current in normal opera- tion is equal to output current. frequency compensation loop frequency compensation is performed on the output of the error amplifier (v c pin) with a series rc network. the main pole is formed by the series capacitor and the output impedance ( ? 500k w ) of the error amplifier. the pole falls in the range of 2hz to 20hz. the series resistor creates a zero at 1khz to 5khz, which improves loop stability and transient response. a second capacitor, typi- cally one-tenth the size of the main compensation capaci- tor, is sometimes used to reduce the switching frequency ripple on the v c pin. v c pin ripple is caused by output voltage ripple attenuated by the output divider and multi- plied by the error amplifier. without the second capacitor, v c pin ripple is: v c pin ripple = v ripple = output ripple (v p? ) g m = error amplifier transconductance ( 1500 mho) r c = series resistor on v c pin v out = dc output voltage 1.245(v ripple )(g m )(r c ) (v out ) to prevent irregular switching, v c pin ripple should be kept below 50mv pCp . worst-case v c pin ripple occurs at maximum output load current and will also be increased if poor quality (high esr) output capacitors are used. the addition of a 0.0047 m f capacitor on the v c pin reduces switching frequency ripple to only a few millivolts. a low value for r c will also reduce v c pin ripple, but loop phase margin may be inadequate. input capacitors the input capacitor of a boost converter is less critical due to the fact that the input current waveform is triangular and does not contain large squarewave currents as is found in the output capacitor. capacitors in the range of 10 m f to 100 m f with an esr of 0.1 w or less work well up to full 6a switch current. higher esr capacitors may be acceptable at low switch currents. input capacitor ripple current for a boost converter is : i ripple = f = 500khz switching frequency 0.3(v in )(v out ?v in ) (f)(l)(v out ) the input capacitor can see a very high surge current when a battery or high capacitance source is connected live and solid tantalum capacitors can fail under this condition. several manufacturers have developed tantalum capaci- tors specially tested for surge capability (avx tps series, for instance) but even these units may fail if the input voltage approaches the maximum voltage rating of the capacitor during a high surge. avx recommends derating capacitor voltage by 2:1 for high surge applications. ceramic, os-con and aluminum electrolytic capacitors may also be used and have a high tolerance to turn-on surges. ceramic capacitors higher value, lower cost ceramic capacitors are now becoming available in smaller case sizes. these are tempt- ing for switching regulator use because of their very low esr. unfortunately, the esr is so low that it can cause loop stability problems. solid tantalum capacitor esr generates a loop zero at 5khz to 50khz that is instru- mental in giving acceptable loop phase margin. ceramic capacitors remain capacitive to beyond 300khz and usu- ally resonate with their esl before esr becomes effective. they are appropriate for input bypassing because of their high ripple current ratings and tolerance of turn-on surges.
11 LT1370 applicatio s i for atio uu w u layout considerations for maximum efficiency, LT1370 switch rise and fall times are made as short as possible. to prevent radiation and high frequency resonance problems, proper layout of the components connected to the switch node is essential. b field (magnetic) radiation is minimized by keeping output diode, switch pin and output bypass capacitor leads as short as possible. figure 3 shows recommended posi- tions for these components. e field radiation is kept low by minimizing the length and area of all traces connected to the switch pin. a ground plane should always be used under the switcher circuitry to prevent interplane coupling. the high speed switching current path is shown schemati- cally in figure 4. minimum lead length in this path is essential to ensure clean switching and low emi. the path including the switch, output diode and output capacitor is the only one containing nanosecond rise and fall times. keep this path as short as possible. more help for more detailed information on switching regulator circuits, please see application note 19. linear technology also offers a computer software program, switchercad tm , to assist in designing switching convert- ers. in addition, our applications department is always ready to lend a helping hand. figure 4 load v out l1 switch node LT1370 ?f04 v in high frequency circulating path v in nfb gnd fb v sw v c s/s c d keep path from v sw , output diode, output capacitors and ground return as short as possible c LT1370 ?f03 figure 3. layout considerations r package switchercad is a trademark of linear technology corporation.
12 LT1370 typical applicatio n s n u positive-to-negative converter with direct feedback LT1370 v in v c v in 2.7v to 13v *bh electronics 501-0726 gnd nfb LT1370 ?ta03 v sw s/s d2 p6ke-15a d3 1n4148 d1 mbrd835l c1 100 m f c2 0.047 m f c3 0.0047 m f r1 2k r3 2.49k 1% r2 2.49k 1% ? out ? ?v c4 100 m f 2 on off v in 3v 5v 9v i out 1.75a 2.25a 3a 2 1 4 t1* 3 ? max i out + + dual output flyback converter with overvoltage protection LT1370 v in fb v c v in 2.7v to 10v *dale lpe-5047-100mb gnd nfb LT1370 ?ta04 v sw s/s p6ke-20a 1n4148 mbrs360t3 mbrs360t3 c1 22 m f r2 6.19k 1% r1 13k 1% c2 0.047 m f c3 0.0047 m f r3 2k r5 2.49k 1% r4 12.1k 1% ? out ?5v v out 15v c4 47 m f c5 47 m f on off 2, 3 8, 9 7 t1* 4 10 1 + + +
13 LT1370 typical applicatio n s n u two li-ion cells to 5v sepic converter** LT1370 v in gnd v in 4v to 9v v c fb LT1370 ?ta05 v sw s/s c1 33 m f 20v c4 0.047 m f c5 0.0047 m f r1 2k r3 6.19k 1% r2 18.7k 1% v out ? 5v c3 100 m f 10v 2 on off l1a* 6.8 m h l1b* 6.8 m h c2 4.7 f c1 = avx tpsd 336m020r0200 c2 = tokin 1e475zy5u-c304 c3 = avx tpsd107m010r0100 bh electronics 501-0726 input voltage may be greater or less than output voltage d1 mbrd835l v in 4v 5v 7v 9v i out 2a 2.2a 2.6a 2.8a ? max i out * ** + + single li-ion cell to 5v LT1370 v in v c gnd fb LT1370 ?ta06 v sw s/s l1* c1** 100 m f 10v single li-ion cell c4** 100 f 10v 2 c2 0.047 m f c3 0.0047 m f r3 2k r2 6.19k 1% r1 18.7k 1% v out ? 5v d1 mbrd835l on off * ** coilcraft do3316p-472 avx tpsd107m010r0100 + + + v in 2.7v 3.3v 3.6v i out 2.5a 3a 3.3a ? max i out
14 LT1370 typical applicatio n s n u laser power supply laser 190 w 1% 1n4002 (all) 0.1 m f 10k v in 10 m f v c v in fb gnd 2.2 m f v in 12v to 25v 150 w mur405 l2 82 m h LT1370 l1 5 4 1 3 2 8 11 hv diodes 1800pf 10kv 0.01 m f 5kv 1800pf 10kv 47k 5w 2.2 m f 0.47 m f l1 = l2 = q1, q2 = 0.47 m f = hv diodes = laser = coiltronics ctx02-11128 gowanda ga40-822k zetex ztx849 wima 3x 0.15 m f type mkp-20 semtech-fm-50 hughes 3121h-p 10k LT1370 ?ta07 v sw q1 q2 + + + coiltronics (407) 241-7876
15 LT1370 package descriptio n u dimensions in inches (millimeters) unless otherwise noted. r package 7-lead plastic dd pak (ltc dwg # 05-08-1462) r (dd7) 0396 0.026 ?0.036 (0.660 ?0.914) 0.143 +0.012 0.020 () 3.632 +0.305 0.508 0.040 ?0.060 (1.016 ?1.524) 0.013 ?0.023 (0.330 ?0.584) 0.095 ?0.115 (2.413 ?2.921) 0.004 +0.008 0.004 () 0.102 +0.203 0.102 0.050 0.012 (1.270 0.305) 0.059 (1.499) typ 0.045 ?0.055 (1.143 ?1.397) 0.165 ?0.180 (4.191 ?4.572) 0.330 ?0.370 (8.382 ?9.398) 0.060 (1.524) typ 0.390 ?0.415 (9.906 ?10.541) 15 typ 0.300 (7.620) 0.075 (1.905) 0.183 (4.648) 0.060 (1.524) 0.060 (1.524) 0.256 (6.502) bottom view of dd pak hatched area is solder plated copper heat sink t7 package 7-lead plastic to-220 (standard) (ltc dwg # 05-08-1422) information furnished by linear technology corporation is believed to be accurate and reliable. however, no responsibility is assumed for its use. linear technology corporation makes no represen- tation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 0.040 ?0.060 (1.016 ?1.524) 0.026 ?0.036 (0.660 ?0.914) t7 (to-220) (formed) 1197 0.135 ?0.165 (3.429 ?4.191) 0.700 ?0.728 (17.780 ?18.491) 0.045 ?0.055 (1.143 ?1.397) 0.165 ?0.180 (4.191 ?4.572) 0.095 ?0.115 (2.413 ?2.921) 0.013 ?0.023 (0.330 ?0.584) 0.620 (15.75) typ 0.155 ?0.195 (3.937 ?4.953) 0.152 ?0.202 (3.860 ?5.130) 0.260 ?0.320 (6.604 ?8.128) 0.147 ?0.155 (3.734 ?3.937) dia 0.390 ?0.415 (9.906 ?10.541) 0.330 ?0.370 (8.382 ?9.398) 0.460 ?0.500 (11.684 ?12.700) 0.570 ?0.620 (14.478 ?15.748) 0.230 ?0.270 (5.842 ?6.858)
16 LT1370 ? linear technology corporation 1998 1370f lt/tp 0198 4k ? printed in the usa part number description comments lt1171 100khz 2.5a boost switching regulator good for up to v in = 40v ltc ? 1265 12v 1.2a monolithic buck converter converts 5v to 3.3v at 1a with 90% efficiency lt1302 micropower 2a boost converter converts 2v to 5v at 600ma in so-8 packages lt1372 500khz 1.5a boost switching regulator also regulates negative flyback outputs lt1373 low supply current 250khz 1.5a boost switching regulator 90% efficient boost converter with constant frequency lt1374 500khz 4.5a buck switching regulator converts 12v to 3.3v at 2.5a in so-8 package lt1376 500khz 1.5a buck switching regulator steps down from up to 25v using 4.7 m h inductors lt1512 500khz 1.5a sepic battery charger input voltage may be greater or less than battery voltage lt1513 500khz 3a sepic battery charger input voltage may be greater or less than battery voltage related parts linear technology corporation 1630 mccarthy blvd., milpitas, ca 95035-7417 l (408) 432-1900 fax: (408) 434-0507 l telex: 499-3977 l www.linear-tech.com

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